Semiconductor integrated circuit for communication

ABSTRACT

An orthogonal modulating circuit for modulating signals of two oscillation frequencies differing in phase by 90° with transmission data (I and Q) is used in common for a plurality of bands, an LC resonance circuit comprising inductances L and a capacitor C is used as the output load on the orthogonal modulating circuit instead of resistors commonly used according to the prior art, and the values of L or C constituting the resonance circuit are switched over between each other according to the transmission band.

CROSS-REFERENCE TO RELATED APPLICATION

The present application claims priority from Patent Cooperation Treatypatent application PCT/JP02/010715 filed on Oct. 16, 2002, the contentof which is hereby incorporated by reference into this application.

TECHNICAL FIELD

The present invention relates to a technique that can be effectivelyapplied to a semiconductor integrated circuit for communication for usein wireless communication apparatuses of a direct conversion formula,and more particularly to a technique that can be effectively applied toa modulating circuit and a demodulating circuit constituting a cellularphone or the like that can transmit and receive signals of a pluralityof bands.

BACKGROUND ART

For a communication apparatus, such as a cellular phone, reductions notonly in size and weight but also in cost are keenly called for. Thiscost saving can be effectively accomplished by reducing such externalparts as a voltage control oscillator (VCO) and a SAW filter. Accordingto the Global System for Mobile Communications (GSM), which is one ofthe digital mobile phone systems currently used in Europe, the mainstream is an offset PLL formula, which does not use the expensive SAWfilter. On the other hand, there is a direct conversion formulaaccording to which high frequency oscillation signals, which constitutethe carrier, are directly modulated with transmission data. This directconversion formula would permit a further saving in cost because theIF-VCO, which generates oscillation signals of an intermediatefrequency, can be eliminated.

In recent years, cellular phones have come to be required to have a dualband formula capable of handling signals of two frequency bands, such asthe GSM of the 880 to 915 MHz band and the digital cellular system (DCS)of the 1710 to 1785 MHz band, or a triple band formula capable ofhandling signals of the personal communication system (PCS) of the 1850to 1910 MHz band in addition to the GSM and the DCS. Conceivably,cellular phones will be required to be adaptable to even more bands inthe future.

In a semiconductor integrated circuit (hereinafter referred to as amodulating/demodulating LSI) for modulating transmission signals andreception signals for use in such a cellular phone adaptable to aplurality of bands, providing a modulating circuit and a demodulatingcircuit for handling signals of each of the multiple bands would entailthe problem of an increased circuit area and accordingly a higher chipcost. In view of this problem, it is conceivable to have the pluralityof bands to use a modulating circuit and a demodulating circuit incommon. However, an attempt at common use of a modulating circuit and ademodulating circuit would encounter difficulty to satisfy therequirements regarding the output level and noise characteristics.

More specifically, in the transmission circuitry for instance, if aconventional orthogonal modulating circuit having a resistor as the loadelement is used as it is, it will be difficult to keep the level ofnoise leaking into the reception frequency band (C/N ratio) as low asrequired. In the reception circuitry on the other hand, common use ofmixers at a later stage by multiple bands would allow the output signalof the low noise amplifier (LNA) of the selected band to leak into theoutput of the LNAs of unselected bands, resulting in a drop in theoutput level of the LNA of the selected band and accordingly adeterioration in noise figure (NF) when amplification is performed bythe mixer at a subsequent stage.

An object of the present invention is to provide a semiconductorintegrated circuit for communication equipped with a modulating circuitcapable of satisfying the noise characteristic requirement (C/N ratio)when an orthogonal modulating circuit is used in common by a pluralityof bands to be adaptable to those multiple bands in the transmissioncircuitry of a direct up conversion formula with a view to restrainingan increase in chip area.

Another object of the invention is to provide a semiconductor integratedcircuit for communication equipped with a demodulating circuit capableof avoiding a deterioration in noise figure (NF) when a mixer fordemodulating reception signals is used in common by a plurality of bandsto be adaptable to those multiple bands in the reception circuitry of adirect down conversion formula with a view to restraining an increase inchip area.

Still another object of the invention is to provide a semiconductorintegrated circuit for communication capable of communication over aplurality of bands and, moreover, requiring neither a SAW filter nor anIF-VCO for generating oscillation signals of an intermediate frequency,both of which would be external parts, and thereby enabling the numberof constituent parts required to be reduced.

The above-stated and other objects and novel features of the inventionwill become more apparent from the following description in thespecification when taken in conjunction with the accompanying drawings.

DISCLOSURE OF THE INVENTION

Typical aspects of the invention disclosed in the present applicationwill be briefly described below.

Thus according to the invention, in a transmission circuitry of a directup conversion formula, an orthogonal modulating circuit for modulatingsignals of two oscillation frequencies differing in phase by 90° withtransmission data (I and Q) is used in common for a plurality of bands,an LC resonance circuit comprising inductances L and a capacitor C isused as the output load on the orthogonal modulating circuit instead ofresistors commonly used according to the prior art, and the values of Lor C constituting the resonance circuit are switched over between eachother according to the transmission band. Resistors, if used as theoutput load on the orthogonal modulating circuit, would invite a voltagedrop and narrow the dynamic range of the orthogonal modulating circuit,making it difficult to achieve a large output amplitude. By contrast,the use of the LC resonance circuit as the output load would invite novoltage drop and make it possible to achieve a high output level.Therefore, if the absolute quantity of noise is the same, common use ofthe orthogonal modulating circuit by multiple bands will not obstructimprovement of noise characteristic (C/N ratio).

While the resonance point of the LC resonance circuit can be altered byvarying the value of L or of C, but it is simpler in the current processto vary the value of C. Further, as the capacitor C to serve as aconstituent element of the resonance circuit, a gate capacitor betweenthe gate terminal and the source-drain terminal of an insulating gatetype field effect transistor (hereinafter referred to as MOSFET) can beutilized. This would enable a capacitor having large capacitance perunit area to be obtained without increasing the required steps of themanufacturing process and to restrain an increase in chip size.

According to another aspect of the invention under the presentapplication, in a reception circuitry of a direct down conversionformula, wherein data signals are directly demodulated by mixing withreception signals oscillation signals of substantially the samefrequency as that of the carrier of the reception signals' band, a firststage low noise amplifier (LNA) for amplifying reception signals isprovided for each stage. At the same time, mixers serving as thedemodulating circuit are used in common for the plurality of bands, anda buffer which takes on a high output impedance in their unselectedstate is provided between the LNA of each band and the common mixers.

As the means described above can prevent the output signal of the LNA ofthe selected band from leaking into the output of the LNAs of unselectedbands, and thereby preventing inviting a drop in the output level of theLNA of the selected band, a deterioration in noise figure (NF) of thewhole reception circuitry can be avoided.

Preferably, where the low noise amplifier (LNA) of each band is of adifferential type, a switching element for short circuiting can beprovided between its differential input terminals, and the differentialinput terminals of the LNAs of unselected bands can be placed at thesame potential. This would make it possible to prevent reception signalsand noise having infiltrated from the input side of the LNA of theselected band from leaking out to the output side of the LNA of theselected band and thereby preventing deterioration of noisecharacteristics.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 is a block diagram of an example of transmission circuitry, towhich the present invention is applied.

FIG. 2 is a circuit diagram of a first typical configuration of anorthogonal modulating circuit constituting the transmission circuitryembodying the invention.

FIG. 3 is a circuit diagram of a second typical configuration of anorthogonal modulating circuit constituting the transmission circuitryembodying the invention.

FIG. 4 is a graph showing the relationship between the control voltagefor the variable capacitor and the capacitance of a resonance circuit asthe output load of the orthogonal modulating circuit.

FIG. 5 is a graph showing the frequency characteristic of the orthogonalmodulating circuit in the embodiment of the invention.

FIG. 6 is a circuit diagram of a third typical configuration of anorthogonal modulating circuit constituting the transmission circuitryembodying the invention.

FIG. 7 illustrates an example of reception circuitry studied prior tothe present invention and problems involved in it.

FIG. 8 is a block diagram of a first example of reception circuitry towhich the invention is applied.

FIG. 9 is a circuit diagram of a typical example of configuration of abuffer circuit constituting the reception circuitry in the embodiment.

FIG. 10 is a circuit diagram of a typical example of configuration of alow noise amplifier (LNA) constituting the reception circuitry in theembodiment.

FIG. 11 is a block diagram of a second example of reception circuitry towhich the invention is applied.

FIG. 12 is a block diagram of one example of wireless communicationsystem using a modulating/demodulating LSI pertaining to the invention.

FIG. 13 is a block diagram of schematic configuration of a cellularphone as an example of wireless communication system using themodulating/demodulating LSI pertaining to the invention.

BEST MODE FOR CARRYING OUT THE INVENTION

Next will be described preferred embodiments of the present inventionwith reference to the accompanying drawings.

FIG. 1 shows a first preferred embodiment of the invention, which is atransmission circuitry of a direct up conversion formula.

In FIG. 1, reference numeral 100 denotes a semiconductor integratedcircuit for communication use (hereinafter referred to as amodulating/demodulating LSI) in the transmission circuitry embodying theinvention; 200, a baseband circuit; and 300, a local voltage controloscillator (VCO) generating a local oscillation signal. The basebandcircuit 200 converts transmission data into I signals and Q signals andsupplies them to the modulating/demodulating LSI 100, and generatescontrol signals for the modulating/demodulating LSI 100. It consists ofone or a few semiconductor integrated circuits (IC). The I and Q signalsmay be, though not necessarily are, supplied as differential signals I,/I, Q and /Q to the modulating/demodulating LSI 100. The local VCO 300can generate a local oscillation signal φvco of 3.42 to 3.98 GHz infrequency, and oscillates at a frequency matching the selected band inresponse to a switch-over signal from the baseband circuit 200.

The transmission circuitry comprises attenuators 11A and 11B forconverting I and Q signals from the baseband circuit 200 into signals ofa prescribed level, low-pass filters 12A and 12B for clearing the I andQ signals of high frequency noise, a frequency dividing phase shiftercircuit 13 for dividing the frequency of the local oscillation signalφvco from the local VCO 300 by ½ and generating oscillation signals φ1and φ0 differing in phase from each other by 90°, an orthogonalmodulating circuit 14 for subjecting the oscillation signals φ1 and φ0to orthogonal modulation with the I and Q signals, an amplifier circuit15A connected to an output terminal for GSM, an amplifier circuit 15Bconnected to an output terminal for DCS1800/PCS1900, and a controlcircuit 16 for controlling the inside of the modulating/demodulatingLSI. Though not shown, a reception circuitry is also provided within themodulating/demodulating LSI 100.

The levels of attenuation by the attenuators 11A and 11B are controlledwith control signals from the control circuit 16. Since the outputlevels of the I and Q signals differ in some of the baseband circuitscurrently available for practical use, the attenuation levels of theattenuators 11A and 11B are so adjusted that the levels of the I and Qsignals supplied to the orthogonal modulating circuit 14 be constantirrespective of whatever baseband circuit may be used. This adjustmentof the attenuation levels of the attenuators 11A and 11B is accomplishedin accordance with a value set in advance by the baseband circuit 200 toa register CRG in the control circuit 16. Into the register CRG in thecontrol circuit 16 is also set by the baseband circuit 200 informationdesignating the band to be used.

As a frequency dividing circuit DVD for dividing the frequency of theoscillation signal φvco by ½ and a change-over switch SW0 for changingover the frequency-divided signal or the signal before the frequencydivision are provided in the modulating/demodulating LSI 100, theoscillation signal φvco of the local VCO 300 is used as a commonoscillation signal for GSM, DCS1800 and PCS1900. Thus, a signalresulting from ½ frequency division of the oscillation signal φvco bythe frequency dividing circuit DVD is used for GSM, and the oscillationsignal before the frequency division is used for DCS1800/PCS1900.

The change-over switch SW0 performs changing over in accordance with acontrol signal from the control circuit 16. The control circuit 16supplies a signal to switch over the change-over switch SW0 on the basisof information on band designation set in a register. Changing over theoscillation frequency of the local VCO 300 at an instruction from thebaseband circuit 200 according to the band to be used causes thetransmission frequency to be switched over.

The orthogonal modulating circuit 14 comprises a mixer MIXa formixingthe I signal from the baseband circuit 200 and oscillation signals φ0and /φ0 from the frequency dividing phase shifter circuit 13, a mixerMIXb for mixing the Q signal from the baseband circuit 200 andoscillation signals φ1 and /φ1 from the frequency dividing phase shiftercircuit 13, and a common output load COL on these mixers.

Further in this embodiment of the invention, the orthogonal modulatingcircuit 14 is disposed to be commonly used for each band, and the outputof the orthogonal modulating circuit 14 is supplied to both theamplifier circuit 15A for GSM and the amplifier circuit 15B forDCS1800/PCS1900. As DCS1800 and PCS1900 are close to each other infrequency band, only one amplifier circuit 15B is shared by the twobands in this embodiment, but one separate amplifier circuit may as wellbe provided for each.

FIG. 2 shows a specific example of configuration of the orthogonalmodulating circuit 14. As shown in FIG. 2, the orthogonal modulatingcircuit 14 of this embodiment, the common output load COL is composed ofa pair inductances L1 and L2 arranged in parallel and a variablecapacitor Cv. The variable capacitor Cv is connected between oneterminal each (nodes n1 and n2) of the inductances L1 and L2, and acontrol voltage Vc1 for controlling the capacitance of this variablecapacitor Cv is generated on the basis of the information on banddesignation set in the aforementioned register in the control circuit16. More specifically, the control voltage Vc1 is so generated that thecapacitance of the variable capacitor Cv be raised when in the GSMtransmission mode and lowered when in the DCS1800 or PCS1900transmission mode.

To add, an arrangement in which a plurality of capacitance elements areprovided in parallel and the capacitance element to be connected isselected with a switch is also conceivable as an alternative to theconfiguration in which the capacitance is changed over by varying thevoltage applied to one terminal of the variable capacitor Cv as in thisembodiment, the design would have to reflect advance consideration forthe resistance component of the selector switch because the resonancepoint would vary according to the relative magnitude of this resistancecomponent. Therefore, it is more advantageous in terms of designing easeto change over the capacitance by varying the voltage Vc1 applied to oneterminal of the variable capacitor Cv as in this embodiment.

The mixer MIXa is configured of transistors Q11, Q12, Q21 and Q22 towhose bases are supplied an oscillation signal φ0 from the frequencydividing phase shifter circuit 13 and an oscillation signal /φ0 off theoscillation signal φ0 in phase by 180° and whose emitters are commonlyconnected, a transistor Q13 which is connected to the common emitter ofthe transistors Q11 and Q12 and whose base terminal is supplied with theI signal from the baseband circuit 200, a transistor Q23 which isconnected to the common emitter of the transistors Q21 and Q22 and whosebase terminal is supplied with a reversed-phase signal /I to the Isignal from the baseband circuit 200, and emitter resistances R1 and R2of the transistors Q13 and Q23. The collector terminals of thetransistors Q11 and Q21 are connected to the connecting node n1 for theinductance L1 and the variable capacitor Cv of the common output loadCOL, and the collector terminals of the transistors Q12 and Q22 areconnected to the connecting node n2 for the inductance L2 and thevariable capacitor Cv.

The mixer MIXb is configured of transistors Q31, Q32, Q41 and Q42 towhose bases are supplied an oscillation signal φ1 from the frequencydividing phase shifter circuit 13 and an oscillation signal /φ1 off theoscillation signal φ1 in phase by 180° and whose emitters are commonlyconnected, a transistor Q33 which is connected to the common emitter ofthe transistors Q31 and Q32 and whose base terminal is supplied with theQ signal from the baseband circuit 200, a transistor Q43 which isconnected to the common emitter of the transistors Q41 and Q42 and whosebase terminal is supplied with a reversed-phase signal /Q to the Qsignal from the baseband circuit 200, and emitter resistances R3 and R4of the transistors Q33 and Q43. The collector terminals of thetransistors Q31 and Q41 are connected to the connecting node n2 for theinductance L2 and the variable capacitor Cv of the common output loadCOL, and the collector terminals of the transistors Q32 and Q42 areconnected to the connecting node n1 for the inductance L1 and thevariable capacitor Cv.

FIG. 3 is a circuit diagram showing in a more specific way the variablecapacitor Cv of the orthogonal modulating circuit 14 shown in FIG. 2.The orthogonal modulating circuit 14 in this embodiment uses the gatecapacities of MOSFETs Qc1 and Qc2 as the variable capacitor Cvconstituting part of the common output load COL. Where themodulating/demodulating LSI in this embodiment has, as an internalcircuit, what consists of a so-called Bi-CMOS circuit composed of abipolar transistor and MOSFETs, the variable capacitor Cv can be formedwithout increasing the required steps of the manufacturing process byusing the gate capacities of the MOSFETs as the variable capacitor Cv asstated above. Furthermore, as the gate capacities of MOSFETs arerelatively large in capacitance per unit area, a substantial increase inchip size can be restrained even where the inductances L1 and L2 and thevariable capacitor Cv are provided in place of the resistors inconventional mixers.

FIG. 4 shows the relationship between the capacitance of the variablecapacitor Cv and the control voltage for controlling it. In FIG. 4, abroken line A represents the variation of the capacitance in a casewherein the variable capacitor Cv is composed of one MOSFET whose W/Lratio (the ratio between the gate width and the gate length) is 16 μm/2μm, and a solid line B, a case wherein it is composed of 24 MOSFETswhose W/L ratio is 16 μm/2 μm. If the capacitance per MOSFET when avoltage of 1.5 V or above is applied to the gate is 0.125 pF, a totalcapacitance of 3 pF will be obtained where 24 MOSFETs are connected inparallel. Therefore, the number of MOSFETs to constitute the commonoutput load COL or the gate width can be set according to thecapacitance required by the common output load COL.

FIG. 5 shows the frequency characteristic of the orthogonal modulatingcircuit 14 of FIG. 2 when the capacitance of the variable capacitor Cvand the control voltage Vc1 for controlling it are switched over to 0 Vand 2.8 V. In FIG. 5, a curve a represents the characteristic where 2.8V is applied as the control voltage Vc1, and a curve b, where 0 V isapplied as the control voltage Vc1 . The frequencies fo2 and fo1 at thepeaks of the respective curves are the resonance points of the commonoutput load COL consisting of an LC resonance circuit. In thisembodiment, the capacitance of the variable capacitor Cv is so set thatthese frequencies fo2 and fo1 come close to the 900 MHz of GSM and 1800MHz of DCS/PCS, respectively. To add, a broken curve c in FIG. 5represents the frequency characteristic of the orthogonal modulatingcircuit 14 wherein a resistor is used as the common output load COL. Bycomparing these characteristics, it will be seen that the gain of theorthogonal modulating circuit 14 can be increased in the vicinities ofthe desired frequencies fo2 and fo1.

FIG. 6 shows another example of circuitry for the orthogonal modulatingcircuit 14 of FIG. 2. The orthogonal modulating circuit 14 in thisexample has three pairs of MOSFETs M11/M12, M21/M22 and M31/M32 as thevariable capacitor Cv constituting the common output load COL, and isprovided with separate control voltages Vc1, Vc2 and Vc3 to be appliedto the source-drain terminals of each pair of MOSFETs so that thecapacitance of the variable capacitor Cv consisting of the gatecapacitances of the MOSFETs can be varied by eight steps by changingover these control voltages Vc1, Vc2 and Vc3. Table 1 shows therelationship between the control voltages Vc1, Vc2 and Vc3 and thecapacitance of the variable capacitor Cv. TABLE 1 Vc3 Vc2 Vc1Capacitance of variable capacitor L L L 0 L L H Cva L H L Cvb L H HCva + Cvb H L L Cvc H L H Cva + Cvc H H L Cvb + Cvc H H H Cva + Cvb +Cvc

In Table 1, Cva represents the gate capacitance of the MOSFETs M11 andM12 when the control voltage Vc1 to be applied to the source-drainterminals of the MOSFETs M11 and M12 is set to a high level “H”, such as2.8 V; Cvb, the gate capacitance of the MOSFETs M21 and M22 when thecontrol voltage Vc2 Vc1 to be applied to the source-drain terminal ofthe source-drain terminals of the MOSFETs M21 and M22 is set to the highlevel “H”; and Cvc, the gate capacitance of the MOSFETs M31 and M32 whenthe control voltage Vc3 to be applied to the source-drain terminal ofthe MOSFETs M31 and M32 is set to the high level “H”. A frequencycharacteristic appropriate for the band to be used can be provided tothe orthogonal modulating circuit 14 by so switching over the controlvoltages Vc1, Vc2 and Vc3 as to select the most suitable one out ofthese capacitances according to the frequency band to be used.

Next will be described an example of reception circuitry to which thepresent invention is applied.

In configuring a reception circuitry adaptable to dual bands or triplebands, a mixer for mixing reception signals and local oscillationsignals is usually provided for each individual band. However, thisentails enlarged circuit dimensions and an increased chip size. In viewof this problem, the present inventors considered arrangement of a lownoise amplifier (LNA) for each individual band, as represented by 17Athrough 17C in FIG. 7, and sharing of mixers MIX1 and MIX2 andprogrammable gain amplifiers (PGAs) 18A and 18B at subsequent stages bythe plurality of bands.

As a result, for instance in a case wherein reception signals from anantenna AT are input the LNA 17A via a change-over switch SWc accordingto the selected band, part of the output of the selected LNA 17A leaksinto the other unselected LNAs 17B and 17C to invite a drop in theoutput level of the selected LNA 17A. It was found out that theamplification of this reduced output of the LNA 17A by the mixers MIX1and MIX2 would entail a deterioration in NF in the whole receptioncircuitry. It was also discovered that, since the frequency of receptionsignals differs from band to band, the extent of the LNA output dropalso differs from band to band, and the NF value is varied by a bandswitch-over.

FIG. 8 shows an example of reception circuitry of the direct downconversion formula which embodies the invention. It differs from thecircuitry of FIG. 7 in that buffers 19A, 19B and 19C whose outputimpedances are high when not selected are provided at a stage subsequentto the LNA 17A through LNA 17C, and the outputs of these buffers 19Athrough 19C are input the common mixers MIX1 and MIX2.

Incidentally, since the mixers MIX1 and MIX2 mix the reception signalswith the oscillation signals φ1 and φ0 differing in phase from eachother by 90°, supplied from the frequency dividing phase shifter circuit13′, the I signals and Q signals are thereby demodulated. Though notshown, a variable gain amplifier is provided at a stage preceding themixers MIX1 and MIX2.

Though not limited to this arrangement, the band change-over switch SWcis so switched over that the LNA 17A amplify signals of PCS's 1900 MHzband, the LNA 17B amplify signals of DCS's 1800 MHz band, and the LNA17C amplify signals of GSM's 900 MHz band.

FIG. 9 shows in a more specific way the circuitry of the buffers 19A,19B and 19C at a stage subsequent to the LNAs 17A, 17B and 17C for thedifferent bands constituting the reception circuitry of FIG. 8. Sincethe buffers 19A, 19B and 19C have the same configuration, the followingdescription of the configuration will focus on the buffer 19A.

As shown in FIG. 9, the buffer 19A (19B or 19C) is configured ofcollector-grounded type transistors Q51, Q52, Q61 and Q62 to whose basesthe differential output of the LNA 17A (17B or 17C) are applied viacapacitors Cd1 and Cd2 which cut the respective D.C. components and abiasing and selecting circuit 190 which provides base bias voltages tothese transistors and selectively places one of them in an operatingstate according to the band that is to be used.

Between the transistors Q51 and Q52 of each of the buffers, the emittersare coupled to each other, connected to a grounding point via commonemitter resistors Re1 and Re2. Between the transistors Q61 and Q62 ofeach buffer as well, the emitters are coupled to each other, connectedto a grounding point via common emitter resistors Re3 and Re4. Thepotentials of the common emitters of the transistors Q51 and Q52 areinput to the mixer MIX1. On the other hand, the potentials of the commonemitters of the transistors Q61 and Q62 are input to the mixer MIX2.

The biasing and selecting circuit 190 is configured of resistors Rb1 andRb2 connected to the bases of the transistors Q51 and Q52 of each of thebuffers, a P-channel MOSFET MP1 connected between the connecting nodefor these resistors Rb1 and Rb2 and a source voltage terminal Vcc, andN-channel MOSFETs MN1 and MN2 whose drains are connected to the bases ofthe transistors Q51 and Q52 and each of the buffers. The sources of theN-channel MOSFETs MN1 and MN2 are coupled to each other, and to thesecommon sources is applied a constant voltage Vcs set to be lower thanthe emitter voltage of the emitter follower transistors Q51 and Q52.

The P-channel MOSFET MP1 and the N-channel MOSFETs MN1 and MN2 aresubjected to complementary on/off control by the output of a CMOSinverter INV which inverts selection signals SEA, SEB and SEC for therespectively matching LNAs 17A, 17B and 17C. The selection signals SEA,SEB and SEC are generated and supplied by the control circuit 16 on thebasis of a value set in the register in the control circuit 16 of FIG. 2from the baseband circuit.

Next will be described the actions of the buffers 19A through 19C.Supposed here is a case in which the LNA 17A is selected and the LNAs17B and 17C are not. In this case, the selection signal SEA is set to ahigh level, and the signals SEB and SEC, to a low level. In the biasingand selecting circuit 190 at a stage subsequent to the selected LNA, theP-channel MOSFET MP1 is thereby turned on and the N-channel MOSFETs MN1and MN2 turned off. This causes a D.C. bias potential close to thesource voltage Vcc to be applied to the bases of the transistors Q51,Q52, Q61 and Q62, and the A.C. output of the LNA 17A (reception signal)is superposed over it to be inputted. As a result, an A.C. voltagematching the output of the LNA 17A is generated at the emitters of thetransistors Q51 through Q62 of the buffer 19A for the selected band.

On the other hand, in the biasing and selecting circuits 190 of thebuffers 19B and 19C at a stage subsequent to the LNAs 17B and 17C forunselected bands, the P-channel MOSFET MP1 is turned off and theN-channel MOSFETs MN1 and MN2 are turned on by the selection signals SEBand SEC. This causes a D.C. bias potential from a constant voltagesource Vcs to the bases of the transistors Q51, Q52, Q61 and Q62 to turnoff these transistors Q51 through Q62. In this state, even if the LNAfor the selected band amplifies the reception signals to cause currentsto flow to the emitter follower transistors Q51 through Q62 of thebuffer to cause the emitter potential to vary, the variation isprevented from being conveyed to the output terminals of the inactiveLNAs for the unselected bands. In other words, the output impedances ofthe unselected LNAs are raised. As a result, the level of the outputfrom the LNA of the selected band to the mixers MIX1 and MIX2 isprevented from falling, and nor will there be a deterioration in NFvalue.

FIG. 10 shows another example of reception circuitry which embodies theinvention.

In this embodiment, short-circuiting MOSFETs Ms1, Ms2 and Ms3 areprovided between the base terminals of transistors Q71 and Q72constituting LNAs 17A, 17B abd 17C for multiple bands, and these MOSFETsMs1, Ms2 and Ms3 are controlled with the selection signals SEA, SEB andSEC used for controlling the biasing and selecting circuit 190 in theabove-described embodiment. At a stage subsequent to the LNAs 17A, 17Band 17C, the buffers 19A, 19B and 19C are provided as in the earlierembodiment.

In this embodiment, control is so effected that the short-circuitingMOSFET Ms of the LNA for the selected band be turned off, and theshort-circuiting MOSFETs Ms of the LNAs for the unselected bands beturned on. While this causes the LNA for the selected band to engage inusual differential amplification, the LNAs for the unselected bands, astheir differential input terminals are short-circuited between eachother, are prevented from letting reception signals or noise havingfound their way into the input terminals of the LNAs for the unselectedband be input to the mixers shared by the multiple bands.

Where the configuration is such that the output impedances of unselectedLNAs are raised by providing the buffers 19A, 19B and 19C to improve theNF characteristic of the front end section of the reception circuitry asin the example shown in FIG. 8, the output nodes of the unselected LNAsare placed in a floating state and, conversely, made susceptible to theinfluence of noise from the input side, which might conceivably be inputto the mixers via parasitic capacitors. However, the presence of theshort-circuiting MOSFETs Ms on the input side of LNAs, which are turnedon, can prevent noise having found its way into the LNAs for theunselected bands from being amplified and input into the mixers via thebuffers whose output impedances have been raised. The NF characteristicis further improved as a result.

FIG. 11 shows still another example of reception circuitry whichembodies the invention.

In this embodiment, apart from the LNAs 17A, 17B and 17C for multiplebands, a dummy LNA 17D and a dummy buffer circuit 19D of the samecircuit configuration as their earlier described counterparts areprovided on the input side of the common mixers MIX1 and MIX2. Thesedummy LNA 17D and dummy buffer circuit 19D effectively function wherevariable gain amplifiers (PGAs) 18A and 18B at a subsequent stage areprovided with D.C. offset canceling circuits.

Thus, where the PGAs have offset canceling circuits, usually the LNAs17A, 17B and 17C are placed in a non-operating state when D.C. offsetsare calibrated in the PGAs 18A and 18B. However, since the LNAs areactivated in an actual operation of reception, leaked noise oroscillation signals from the oscillator enter into the mixers via thisactivated selected LNA, the noise having infiltrated via the LNA wouldbe self-mixed by the mixers to give rise of D.C. offsets in the absenceof the dummy LNA 17D and the dummy buffer circuit 19D.

By contrast in this example of circuitry, by virtue of the presence ofthe dummy LNA 17D and the dummy buffer circuit 19D, it is possible toperform calibration in a state in which stray noise resulting fromleaked noise or oscillation signal from the oscillator is given to themixers via the dummy LNA by placing the genuine LNAs 17A, 17B and 17C ina non-operating state and instead performing calibration in the PGAs 18Aand 18B in a state in which the dummy LNA 17D and the dummy buffercircuit 19D are activated. Accordingly, DC offsets are restrained.

Next will be described with reference to FIG. 12 a typical configurationof a mobile communication system using a triple band typemodulating/demodulating LSI using the transmission circuitry and thereception circuitry according to the present invention. The samecircuits as in FIG. 1 will be assigned respectively the same referencesigns, and duplication in description will be avoided. In FIG. 12, ATdenotes an antenna for transmitting and receiving signal saves; 411 and412 denote high frequency band-pass filters for clearing transmissionsignals of noise, such as a SAW filter or a LC filter; 420, a highfrequency power amplifier circuit (power module) for amplifyingtransmission signals; 431 and 432, low-pass filters for clearingtransmission signals of high frequency noise;

440, a switch for changing over between transmission and reception;

451 through 453, high frequency band-pass filters for clearing receptionsignals of unnecessary waves, such as SAW filters; 100, themodulating/demodulating LSI described with reference to the earlierembodiment; 200, a baseband circuit (LSI) for converting transmissiondata into I and Q signals and controlling the modulating/demodulatingLSI 100; 300, a high frequency oscillator for use in both transmissionand reception (RFVCO); and 310, a loop filter constituting a PLL circuittogether with the RFVCO 300.

The modulating/demodulating LSI 100 is provided with an RF synthesizer180 which, together with the local VCO 300 and the loop filter 310,constitutes the PLL circuit. This RF synthesizer 180 consists of a phasecomparator circuit for comparing the phases of the VCO output and of areference signal, a charge pump for generating a voltage matching thephase difference and other elements, and generates an oscillation signalφvco of a high frequency, such as 3.42 to 3.98 GHz. In amodulating/demodulating LSI for use in a multi-band communicationsystem, switching over between transmission and reception isaccomplished by the switching of the oscillation frequency of the PLLcircuit according to the band to be used at an instruction from thebaseband circuit 300. This oscillation signal φvco or a signal resultingfrom its ½ frequency division by the frequency dividing circuit DVD issupplied via a change-over switch SW0 to the frequency dividing phaseshifter circuit 131 on the reception side or the frequency dividingphase shifter circuit 13 on the transmission side. The change-overswitch SW0 is switched over by a control signal from the control circuit16.

The control circuit 16 is provided with a control register CRG, andsetting is performed in this register CRG on the basis of a signal fromthe baseband circuit 200. More specifically, a clock signal CLK forsynchronization, a load enable signal LEN as the control signal and datasignals SDATA are supplied from the baseband circuit 200 to themodulating/demodulating LSI 100, and the mode control circuit 16, whenthe load enable signal LEN is asserted to a valid level, the datasignals SDATA transmitted from the baseband circuit 200 are successivelytaken in synchronously with the clock signal CLK to be set into thecontrol register CRG. The data signals SDATA are serially transmitted,though not absolutely limited to serial transmission.

The control register CRG in the control circuit 16 is provided with bitsincluding but not limited to a control bit for designating the band tobe used in the above-described example of circuitry, and a modeselection bit for designating a reception mode, a transmission mode, anidle mode in which, as when waiting, only some of the circuits operateand most circuits including the oscillator circuit are stopped in asleeping state, and a warm-up mode in which the PLL circuit is actuated.

FIG. 13 is a block diagram showing the overall configuration of acellular phone to which the modulating/demodulating LSI of theabove-described embodiment is applied.

The cellular phone in this example is provided with a liquid crystalpanel 320 as the display unit, an antenna 321 for use in transmissionand reception, a loudspeaker 322 for voice outputting, a microphone 323for voice inputting, a liquid crystal control driver 311 for driving theliquid crystal panel 320 to display images, a voice interface 330 forcausing the microphone 323 and the loudspeaker 322 to perform voiceinputting/outputting, a high frequency interface 340 for performingcellular phone communication by a GSM formula or otherwise via theantenna 321, a digital signal processor (DSP) 351 for processing voicesignals and transmission/reception signals, application specificintegrated circuits (ASICs) 352 providing customized functions (userlogic), a system control device 353 consisting of a microprocessor or amicrocomputer for controlling the whole apparatus including displaycontrol, a memory 360 for storing data and programs, and an oscillatorcircuit (OSC) 370 among other elements. The DSP 351, the ASICs 352 andthe microcomputer 353 as the system control device constitute thebaseband circuit 200. The modulating/demodulating LSI in this embodimentis used as the transmitter/receiver unit for the high frequencyinterface 340. In the high frequency interface 340 are arranged a highfrequency power amplifier (power module) and other elements in additionto the modulating/demodulating LSI 100.

While the invention achieved by the present invention has been hithertodescribed with reference to specific embodiments thereof, the inventionis not limited to these embodiments. In the described embodiments, forinstance, both the transmission circuitry and the reception circuitryare supposed to operate in a direct conversion formula, but theconfiguration may as well be such that only one of them operates in adirect conversion formula. Further, though the embodiment is supposed touse the circuit for generating oscillation signals (VCO or the like) incommon for the transmission circuitry and for the reception circuitry,separate such circuits can be provided as well. Although the descriptionof the embodiments referred to what uses a modulating/demodulating LSIadaptable to three bands including GSM, DCS 1800 and PCS 1900, theinvention is also applicable to a modulating/demodulating LSI adaptableto two bands including GSM and DCS 1800, and a modulating/demodulatingLSI adaptable to four or more bands.

Further the embodiment described above uses a variable capacitor Cv tovary the capacitance in order to alter the resonance point of theresonance circuit as the output load on the orthogonal modulatingcircuit 14 on the transmission side, it is also possible to alter theresonance point by varying the inductance.

Advantages achieved by some of the most typical aspects of the inventiondisclosed in the present application will be briefly described below.

Thus according to the invention, in the transmission circuitry of adirect up conversion formula, even if orthogonal modulating circuits toadapt to a plurality of bands are shared among multiple bands torestrain an increase in chip area, the requirement for noisecharacteristic (C/N ratio) can be satisfied. Also according to theinvention, in the reception circuitry of a direct down conversionformula, even if low noise amplifiers (LNAs) for amplifying receptionsignals to adapt to a plurality of bands are shared among multiple bandsto restrain an increase in chip area, it is possible to avoiddeterioration in noise figure (NF). Furthermore, communication withsignals of a plurality of frequency bands is possible, and there is noneed for such external parts as a SAW filter and an IF-VCO forgenerating intermediate frequency oscillation signals. This alsocontributes to reducing the number of constituent parts.

Industrial Applicability

Although the foregoing description mainly concerned the application ofthe invention by the present inventors to modulating/demodulatingcircuits for use in wireless communication systems including cellularphones, which constitute the area of utilization underlying theinventive attempt, the invention is not limited to them, but areextensively applicable to modulating/demodulating circuits in general.

1. A semiconductor integrated circuit for communication with a built-intransmission circuitry of a direct up conversion formula provided withan orthogonal modulating circuit using as a carrier signals of twooscillation frequencies differing in phase from each other andperforming orthogonal modulation of this carrier with I signals and Qsignals based on transmission data, wherein an LC resonance circuitcomprised of inductance elements and a capacitance element is providedas the output load on said orthogonal modulating circuit.
 2. Asemiconductor integrated circuit for communication with a built-intransmission circuitry of a direct up conversion formula provided withan orthogonal modulating circuit using as a carrier signals of twooscillation frequencies differing in phase from each other andperforming orthogonal modulation of this carrier with I signals and Qsignals based on transmission data, and performing modulation oftransmission signals of a plurality of frequency bands by changing overthe frequency of said carrier, wherein said orthogonal modulatingcircuit includes as the output load thereon an LC resonance circuitcomprised of inductance elements and a capacitance element, and isprovided with common mixers to perform orthogonal modulation on thecarrier of said plurality of frequency bands, and wherein said LCresonance circuit is configured to allow the resonance point to bevaried by a control signal generated according to the selectedtransmission frequency band.
 3. The semiconductor integrated circuit forcommunication according to claim 2, wherein said LC resonance circuit isconfigured to allow the resonance point to be varied by the variation ofthe capacitance of the capacitance element constituting part of the LCresonance circuit by said control signal.
 4. The semiconductorintegrated circuit for communication according to claim 2, wherein saidcapacitance element utilizes a gate capacitance between the gateterminal and the source-drain terminal of an insulating gate type fieldeffect transistor.
 5. The semiconductor integrated circuit forcommunication according to claim 2, wherein said plurality of frequencybands are the 880 to 915 MHz band in a GSM formula, the 1710 to 1785 MHzband in a DCS formula and the 1850 to 1910 MHz band in a PCS formula,and at a stage subsequent to said modulating circuit are provided afirst amplifier circuit for amplifying transmission signals in the GSMformula and a second amplifier circuit for amplifying transmissionsignals in the DCS formula and the PCS formula.
 6. A semiconductorintegrated circuit for communication with a built-in reception circuitryof a direct down conversion formula provided with mixer circuits fordemodulating I signals and Q signals by synthesizing signals of twooscillation frequencies differing in phase from each other intoreception signals, and performing demodulation of reception signals of aplurality of frequency bands by changing over the frequency of saidoscillation frequency signals, wherein first stage amplifier circuitsfor amplifying reception signals respectively matching said plurality offrequency bands are provided, said mixer circuits are provided at astage subsequent to said first stage amplifier circuits as commoncircuits for reception signals of said plurality of frequency bands, andbuffer circuits whose outputs take on high impedances in theirunselected state while matching to said first stage amplifier circuitsare provided between said first stage amplifier circuits and said mixercircuits.
 7. The semiconductor integrated circuit for communicationaccording to claim 6, wherein said buffer circuits comprise emitterfollowers, each connected to the output terminal of one or another ofsaid amplifier circuits via a capacitance element, and biasing circuitsfor generating the base bias voltages of transistors constituting theemitter followers.
 8. The semiconductor integrated circuit forcommunication according to claim 6, wherein said emitter followers areconfigured to share emitter resistances among one another.
 9. Thesemiconductor integrated circuit for communication according to claim 6,wherein each of amplifier circuits comprises a differential amplifiercircuit having a pair of differential input terminals, and a switchingelement for achieving electrical short-circuiting between said pair ofdifferential input terminals is provided.
 10. The semiconductorintegrated circuit for communication according to claim 6, wherein avariable gain amplifier circuit having an offset canceling circuit isprovided on the output side of said mixer circuits, and a dummyamplifier circuit and a dummy buffer circuit having respectively thesame configurations as said first stage amplifier circuits and buffercircuits but not involved in the amplification of reception signals areprovided on the output side of said mixer circuits, and wherein saidoffset canceling circuit is so configured as to perform offsetcalibration in a state in which the genuine first stage amplifiercircuits for amplifying reception signals and buffer circuits are placedin an inactive state and said dummy amplifier circuit and dummy buffercircuit are in an activated state.
 11. A semiconductor integratedcircuit for communication comprising: a transmission circuitry of adirect up conversion formula provided with an orthogonal modulatingcircuit using as a carrier signals of two oscillation frequenciesdiffering in phase from each other and performing orthogonal modulationof this carrier with I signals and Q signals based on transmission data,and performing modulation of transmission signals of a plurality offrequency bands by changing over the frequency of said carrier; and areception circuitry of a direct down conversion formula provided with aplurality of first stage amplifier circuits for amplifying receptionsignals respectively matching a plurality of frequency bands, and mixercircuits for demodulating I signals and Q signals by synthesizingsignals of two oscillation frequencies differing in phase from eachother and amplified by said first stage amplifier circuits intoreception signals, and performing demodulation of a plurality offrequency bands by changing over the frequency of said oscillationfrequency signals, wherein said orthogonal modulating circuit has as theoutput load thereon an LC resonance circuit comprised of inductanceelements and a capacitance element, and is provided with common mixersto perform orthogonal modulation on the carrier of said plurality offrequency bands, and said LC resonance circuit is configured to allowthe resonance point to be varied by a control signal generated accordingto the selected transmission frequency band, wherein said mixer circuitsare provided as common circuits for reception signals of said pluralityof frequency bands, and buffer circuits whose outputs take on highimpedances in their unselected state while matching to said first stageamplifier circuits are provided between said first stage amplifiercircuits and said mixer circuits, and wherein oscillation frequencysignals generated by a common oscillator circuit are supplied to saidtransmission circuitry and said reception circuitry.